RFID Reader

ABSTRACT

Embodiments of the invention relate to the field of RFED (radio frequency identification). Some particularly preferred embodiments relate to a high-Q, so-called “full duplex” (FDX) RFID Reader. An RFID tag reader, the reader comprising: an electromagnetic (em) field generator for generating an electromagnetic (em) field for modulation by said tag, said modulation comprising modulated load of said em field by said tag; a detector system responsive to fluctuations in strength of said em field at said reader; a negative feedback system responsive to said detector system to provide a control signal for said em field generator for controlling said em field generator to reduce said detected fluctuations; and a demodulator responsive to said control of said em field to demodulate said em field modulation by said tag.

FIELD OF THE INVENTION

The invention broadly relates to the field of RFID (radio frequencyidentification). Some particularly preferred embodiments relate to ahigh-Q, so-called “full duplex” (FDX) RFID Reader.

BACKGROUND OF THE INVENTION

In an RFID system with a passive transponder generally the reader hastwo main functions. Firstly the reader supplies energy to thetransponder through an RF energising field. The transponder picks thisup with an antenna and resonant circuit tuned to the actuationfrequency. Secondly, once the transponder is powered, the reader alsoneeds to communicate with the remote device. These two tasks are quitedifferent in nature and that can translate to conflicting requirementson the reader.

The powering of the transponder is important in determining the readrange of the system. The transponder and reader are often weakly coupledand care must be taken to maximise the energy transfer from reader totransponder. A resonant circuit is generally used to improve theefficiency by recycling energy in the reader antenna. For maximum energytransfer, a high Q reader antenna matched to the resonant frequency ofthe transponder would give optimal efficiency. However, once powered,the system is required to communicate, and high. Q severely limits thecommunication bandwidth achievable. In fact the communication bandwidthis inversely proportional to the resonance Q, therefore any improvementin the powering efficiency through an increased system Q has a directconsequence of reduced communication bandwidth.

In a half duplex system (HDX) the powering cycle and communication areseparated in time. This provides flexibility in separation of the twofunctions of the reader and a number of approaches in the prior art havebeen proposed to improve the efficiency of power transfer while keepingthe appropriate communication bandwidth. The most basic method is to usea separate circuit for the power and communication links, for exampleU.S. Pat. No. 4,550,444 and U.S. Pat. No. 4,912,471, which can thereforebe optimised separately. The drawback of this approach is addedcomplexity and cost associated with separate circuits. This is remediedby the alternative approach taken in U.S. Pat. No. 5,025,492, U.S. Pat.No. 5,374,930, and U.S. Pat. No. 5,541,604, where the same antenna isused for both power transmission and communication. A damping circuit iscoupled into the resonance when the powering cycle is complete such thatthe powering cycle may be carried out optimally with a high Q antenna,switching to lower Q and therefore wider bandwidth for the communicationcycle.

The above solution to the conflicting Q requirements is made possiblebecause powering and reading functions are separated in time. Incontrast a full duplex system (FDX) does not enable such an approach. Inan FDX system the power from the reader is kept on for the duration ofthe read cycle. The transponder does not contain a separate transmitter,but instead communicates with the reader through modulation of the loadon its pickup coil; the load modulation is picked up and interpreted bythe reader. An FDX system can have the advantage of simpler transponderswith lower power requirements.

Under some circumstances it can be advantageous to provide separatepowering and communication antennae. This however has to be done withcare as both antennae are operating simultaneously and mutual couplingcan introduce problems. For example, simply setting up a poweringantenna with high Q and a closely spaced communication antenna withlower Q will not necessarily offer a benefit. The modulation associatedwith the communication can cause ringing in the powering antenna thatconfuses the pickup signal in the communication antenna due to mutualcoupling. The additional drawback of a multi-coil reader is increasedcomplexity and cost.

There is therefore a need for a single antenna FDX reader that hassimultaneously the properties of high Q for efficient power transfer toa transponder and also wide communication bandwidth.

SUMMARY OF THE INVENTION

The invention is set out in the independent claims.

We describe techniques by which a high Q coil may be used for widebandwidth communications, for example with an FXD RFID transponder.Broadly these arise from the observation that the cause of theconventional bandwidth limitation is that the reader antenna is requiredto change state. More specifically, the load modulation by thetransponder gives rise to modulation of the antenna voltage amplitudeand the reading function is carried out by measuring this amplitudemodulation. Any such change in state of a high Q system requires anincreased timescale, limiting the rate of data transfer. In embodimentsthe modulated load of the em field by the tag may comprise modulatedabsorption.

We describe an approach where negative feedback is introduced betweenthe stimulus signal and the resonance amplitude. This feedback acts tokeep the amplitude of the antenna voltage constant. Now load modulationby the transponder is quickly compensated for by a change in the readerto keep the antenna amplitude constant. In this manner the reader canadapt quickly to changes induced by the transponder, in embodimentsavoiding the rate limitation associated with a high Q antenna. Themodulation signal may be monitored indirectly through the change in thereader input into the resonance. Alternatively, if the feedback is lessstrong and the resonance amplitude is allowed to vary by some degreethen the modulation may be monitored either through the reader input orthrough changes in the resonance amplitude. In this latter case thenegative feedback still acts to reduce the amplitude variation and hencecan increase the speed of response of the reader compared with a systemwithout feedback.

In these embodiments, the transponder or tag communicates via resistiveload modulation. When operating at the resonance frequency of thetransponder, this modulation translates to a resistive transformedimpedance, as seen by the reader. Consequently, when feedback is used tokeep the amplitude of the reader antenna constant, this translates to amodulation of the power delivered to the antenna. The power may beconveniently measured to yield the modulation waveform.

Alternatively the transponder or tag may modulate a reactive component(also load modulation) such as a capacitance, for example to move aresonant frequency of the tag (when, again, absorption of an em field bythe tag would change because the frequency would change), or to causesome other effect detectable at the reader. Broadly, the modulatingresult is a detectable impedance change at the reader.

The high Q of the reader antenna can be beneficial in extending the readrange of the system. The high Q translates to a reduced level of powerdelivered to the antenna, for a given resonance amplitude. Thepercentage modulation in the input power caused by the transponder istherefore enhanced, making it easier to pick up at low levels. The useof a high Q reader antenna therefore has the potential to not onlyincrease the efficiency of the reader but also extend the effective readrange. Embodiments of the RFID system we describe may be incorporated inone or more of the following:

An asset tracking RFID system; an identification RFID system of peopleor animals; an animal feeding control RFID system; an automatic vehicleidentification RFID system; for labelling of products in a retailenvironment; for theft protection or bill totalling; and an RFID systemfor storage information, for example on a credit card or a passport.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a schematic of an embodiment of a FDX RFID reader with PWMfeedback.

FIG. 2A is a plot of the resonance voltage as a function of time, wherethe amplitude of the resonance is ramped up. FIG. 2B is thecorresponding plot of reader antenna current.

FIG. 3A shows the steady state waveform of the stimulus voltage. FIGS.3B and 3C show the corresponding plots for the Schmitt trigger A1 inputand output, respectively.

FIG. 4A shows the reader antenna current together with the outputvoltage from the Schmitt trigger A1. FIG. 4B shows the current intocapacitor C10 and FIG. 4C shows the current supply pulse from theSchottky diode D6.

FIG. 5A is a plot of the capacitor C10 voltage. FIG. 5B is a plot of thevoltage applied to the antenna and FIG. 5C is a plot of the currentsupply pulse from the Schottky diode D6.

FIG. 6A is a plot of the resonance voltage together with the storedvoltage on capacitor C12. FIG. 6B is a zoomed graph of FIG. 6A. FIG. 6Cis a plot of the transistor Q1 base voltage.

FIG. 7A is a plot of the modulated transponder current. FIG. 7B is aplot of the resonance voltage and FIG. 7C is a zoomed plot of the same.

FIG. 8A is a plot of the output pulse from the Schmitt trigger A1 forthe two cases of high and low transponder current. FIG. 8B is thecorresponding plot of the current supply pulse from the Schottky diodeD6

FIG. 9 is a schematic of an embodiment of a FDX RFID reader with PWMfeedback and measurement of the input power into the resonance.

FIG. 10A is a plot of the modulated transponder current and FIG. 10B isthe corresponding plot of the power measurement circuit output.

FIG. 11 is a schematic of an embodiment of a FDX RFID reader without PWMfeedback.

FIG. 12A is a plot of the modulated transponder current. FIG. 12B is aplot of the resonance voltage and FIG. 12C is a zoomed plot of the same.

FIG. 13A is a plot of the stimulus voltage for the two cases of high andlow transponder current. FIG. 13B is the corresponding plot of thecurrent supply pulse from the Schottky diode D6

FIG. 14 is a schematic of an embodiment of a FDX RFID reader without PWMfeedback and with measurement of the input power into the resonance.

FIGS. 15A and 1513 are plots of the power measurement circuit outputvoltage with and without PWM feedback, respectively.

FIG. 16 is a schematic of an embodiment of a FDX RFID reader without PWMfeedback and with measurement of the input power into the resonancethrough the maximum transient voltage.

FIG. 17 shows an embodiment comprising a cat flap controlled by an RFIDreader of a sub-dermal transponder.

FIG. 18 shows an embodiment comprising an antenna, lock, and readerelectronics that is mounted onto a standard cat flap.

In FIGS. 2, 7, 10 and 12, although printing artefacts may appear, theareas within the signal envelopes should be solid because of the timescale on which the waveforms are depicted.

DETAILED DESCRIPTION OF THE EMBODIMENTS

FIG. 1 shows a schematic of an embodiment of the invention. Thisembodiment is based on a method that provides a resonant circuit 1responsive to a wide frequency range. Also shown in the figure are PWMpulse generator 2, feedback circuit 3, deadband delay generator 4,stimulus pulse generator 5, FET gate voltage 6, and stimulus voltage 7.

In embodiments the resonant circuit comprises a controllable electricresonator comprising an inductor coupled to a first capacitor to form aresonant circuit, the resonator further comprising a controllableelement, a second capacitor controllable coupled across said firstcapacitor by said controllable element, and a control device to controlsaid controllable element such that a total effective capacitance ofsaid first and second capacitor varies over a duty cycle of anoscillatory signal on said resonator. Preferably the controllableelement comprises a switching element, in particular a FET; preferablythe control device comprises a bias circuit for the FET.

The operation of the resonant circuit is now outlined and the additionalelements are described in subsequent paragraphs.

The antenna comprises 32 turns of 660-strand 46AWG Litz wire, withoverall diameter approximately 20 cm. Around the target operatingfrequency of 125 kHz the antenna has inductance of 300 μH (L1) andeffective series resistance of 0.70 (R1), giving a Q of 340. The antennais placed in series with the capacitor network C1, C2, C3 and an n-typeFET. The capacitive network presents two different net capacitances inseries with the antenna, depending on whether the FET is on or off. Theduty cycle over which the FET is on depends on the oscillation amplitudethrough the associated variation of the FET source potential. Thenatural resonance frequency of the circuit is therefore determined bythe amplitude. A high level of oscillation gives a near 50% duty cycle,whereas an amplitude less than Vth gives a 0% duty cycle. These twoextremes of duty cycle correspond to two extremes of frequency, given bythe following equations:

$f_{50\%} = \frac{1}{\pi ( {\sqrt{L \cdot ( {{C\; 1} + ( {{C\; 2^{- 1}} + {C\; 3^{- 1}}} )^{- 1}} )} + \sqrt{L \cdot ( {{C\; 1} + {C\; 3}} )}} )}$$f_{0\%} = \frac{1}{2\pi \sqrt{L \cdot ( {{C\; 1} + ( {{C\; 2^{- 1}} + {C\; 3^{- 1}}} )^{- 1}} )}}$

The above equations give a frequency range over which the circuit mayresonate in response to a stimulus.

When a negative voltage is placed on the FET gate then a largeramplitude is required to keep the same duty cycle of FET conduction.Therefore a negative gate voltage may be used to increase the amplitudeof oscillation to the required level. In this manner the resonancecircuit block, 1, shown in FIG. 1 may be used in an RFID reader togenerate an interrogation field over a wide frequency band. One benefitof this system is that excitation of an antenna may be achieved at apre-determined frequency, without fine-tuning of an inductor and/or acapacitor. Furthermore, the system is tolerant to some degree ofdetuning e.g. with metallic or magnetic material placed in the vicinityof the antenna, provided the operating frequency band still encompassesthe stimulus frequency.

FIGS. 2A and 2B show the resonance voltage (FET1 drain) and inductorcurrent as a function of time. Initially the FET gate voltage Vgate, 6,is set at 0V for a period of 1 ms, then decreasing from 0V to −25V overthe next 4 ms. With the decrease in Vgate, the resonance voltageincreases in amplitude, reaching +65V, −39V at 5 ms, then slowlyincreasing to +71V, −41V by 20 ms. The current shows a correspondingincrease to +/−230 mA and then slowly to 253 mA. The further slowincrease in the amplitude is linked to the feedback circuit and isdescribed later. The gate voltage controlling the resonance amplitude(Vgate, 6) is shown as an ideal voltage source. Because the gate voltagerequired may exceed the available voltage rails, for example from abattery, the control circuitry for Vgate may use a negative voltage railgenerated from the resonance. Such a negative rail builds up inmagnitude with the resonance, giving sufficient level to set the gatevoltage.

The pulse train applied to the stimulus FET pair (FET3 and FET4) isgenerated with pulse width modulation (PWM). Changes in the resonanceamplitude are fed back to determine the pulse width. The steady stateresponse is now used to illustrate the intended operation of thecircuit. First the PWM generator, 2, waveforms are shown in FIG. 3. FIG.3A shows the Vstimulus, 7, waveform controlled by an external sourcesuch as a microcontroller. In this example the stimulus waveform is a 2μs square pulse repeated every 8 μs i.e. 125 kHz repeat rate. When thestimulus voltage is positive it turns on FET2, which zeros the input toSchmitt trigger A1. This is clear in the voltage waveform of the inputto A1 shown in FIG. 3B. When the stimulus voltage is low FET2 is off andthe A1 input increases through charging of C13 by the transistor Q1. Thespeed of this charging is controlled by the 10 k series resistor R10.The asymptote of the charging is 3.5V, 0.6V below the transistor basevoltage of 4.1V. When the input to A1 exceeds the transition thresholdof the Schmitt trigger then the output switches, as shown in FIG. 3C.The overall pulse width generated is approximately 3.5 μs, in partdetermined by the stimulus pulse and in part by the voltage ramp. Notethat this discussion ignores the feedback path from the resonancevoltage amplitude to the transistor base, which will be consideredbelow.

The 3.5 μs pulse is passed on to the deadband delay generator, 4, thatinverts the pulse and introduces a deadband delay to minimise any shootthrough current in the complementary FET pair FET3/FET4. The p-type FET3is conducting for the duration of the 3.5 μs pulse and the n-type FET4for the remainder of the 8 μs cycle. FIG. 4A shows the p-type FET3 gatevoltage pulse (in arbitrary units) together with the inductor current.It is clear that the two waveforms are approximately 90 degrees out ofphase, therefore most of the current through FET3 is transient, drawinglittle net power from the supply. When FET3 turns on, the current intothe FET source is negative, flowing out of the FET into C10. This raisesthe voltage on C10 above its steady state level for the duration of thetransient current, as shown in FIG. 5A, with the diode D6 cutting thecircuit off from the 5V power. The transient current flows first intoC10 and then back out, as shown in FIG. 4B. When zero net current hasflown, any further current lowers the potential of C10 such that theSchottky diode D6 starts to conduct. This diode provides a current pulsefrom the 5V power supply that keeps the resonance running. FIG. 5B showsthe corresponding waveform of the voltage input to the resonance circuitand FIG. 5C shows the current pulse from D6 for comparison. Thisarrangement therefore supplies through D6 only enough current to keepthe resonance amplitude constant.

The feedback circuit, 3, is now considered. FIG. 6A shows the resonancevoltage together with the voltage stored on C11, the other side of thediode D7. FIG. 6B shows the same graph zoomed in. The stored voltage hasan approximately sawtooth waveform where it decays by about 0.26V due tothe 4.7M resistor placed across C11, and is topped up by the resonancevoltage every cycle. The voltage stored on C11 is thereforerepresentative of the resonance amplitude and is also sensitive to smallchanges in both directions (up to 0.26V per 8 μs cycle). In fact,because of the direction of the diode D7, any larger changes ofincreasing amplitude will be transferred to the stored voltage, howeverthe decay rate set by R7 determines the sensitivity for changes to loweramplitude. This voltage is high pass filtered when it is passed throughC12 to the base of the transistor. The filter frequency is set by C12and the input impedance of the transistor amplifier, which is dominatedby the bias resistors R8 and R9.

In the steady state, the resonance amplitude is fixed and the transistorbase voltage, shown in FIG. 6C, sits at 4.1V, set by the bias resistors.There is also a small ripple that is passed on to the base through thehigh pass filter of C12. These together set the pulse with, whichremains constant. Now considering a change in resonance amplitude due toa transponder modulating, any transient increase or decrease in theresonance amplitude is passed on to the transistor base. If theresonance amplitude increases then this increases the transistor basevoltage, increasing the current that charges C13 and therefore reducingthe pulse width. This reduction in pulse width reduces the energy intothe resonance, acting to lower the resonance amplitude. The reverseeffect takes place with a transient reduction in the resonance amplitudetherefore negative feedback has been implemented to keep the amplitudeconstant. At this point it is noted that this negative feedback isresponsible for the later slow ramp of the resonance amplitudeidentified in FIG. 2. The low filter frequency is set by C12 and theinput impedance of the transistor amplifier controls the time constantof this slow ramp.

In order to demonstrate the effectiveness of the feedback, a transponderis introduced into the circuit. The transponder comprises a 1 mHinductor, with parallel capacitance 1.6 nF giving a resonant frequencyof 125 kHz. The Q of the transponder, set by the 50Ω effective seriesresistance, is approximately 15. The coupling to the reader antenna isset to 1%. FIG. 7A shows the transponder current, where the modulationis clear. FIG. 7B shows the corresponding antenna resonance voltage withvery little visible variation. The level of variation is clearer in FIG.7C, where the zoomed in graph shows a peak-to-peak amplitude variationof 150 mV. This is amplitude is very low, demonstrating the efficacy ofthe negative feedback to keep the resonance constant amplitude.Furthermore, the timing of the transponder current waveform is wellrepresented in the residual variation of the antenna voltage,demonstrating that the reader is responding quickly to the transponder,substantially unlimited by high Q rate constraints.

FIG. 8 illustrates the change in input power by the reader in responseto the transponder modulation. FIG. 8A shows the output of the Schmitttrigger A1 in the neighbourhood of the times 300 μs and 500 μs,corresponding to high and low current in the transponder, respectively.When the transponder current is high the effect of the feedback is toincrease the width of the PWM, thus compensating for the additionalenergy dissipated in the transponder. The current pulses supplied bySchottky diode D6 are shown in FIG. 8B, where the high current situationcorresponds to a significant widening, particularly towards later time.This widening of the PWM pulse is responsible for the additional energysupplied to the resonance. As mentioned earlier, the proportional effecton the total energy supplied is amplified up due to the high Q readerantenna.

The total energy supplied to the resonance is therefore a sensitivemeasure of the transponder modulation. FIG. 9 shows a schematic of thefirst embodiment with an additional section that measures current pulsesthough D6. A current measurement resistor R11 generates voltage spikesin response the diode current, which are capacitively coupled to rampgenerator Q2. This generates a sawtooth waveform where the maximumvoltage is mostly controlled by the duration of the current supplypulse. The maximum voltage is stored on C14 (less the Vbe drop of Q3),which also includes a decay resistor R16 such that this voltage maypickup both increasing and decreasing variations. Lastly Q4 providesgain and low pass filtering to reduce the amplitude of the rippleassociated with the R16.

FIG. 10A shows the modulated transponder current and FIG. 10B shows theaccompanying output voltage waveform, Vout. Vout provides an excellentmeasure of the modulation, especially exhibiting the different rates atwhich the transponder current builds up and is cut off. The reader doesnot add significantly to the time constants set by the Q of thetransponder coil only, demonstrating how negative feedback has beenemployed to escape rate constraints associated with a high Q readercoil. The output waveform shown in FIG. 10B may be passed on to a levelswitch such as a Schmitt trigger or alternatively an analogue to digitalconverter (ADC) for subsequent interpretation of the transpondermodulation signal.

FIG. 11 shows an alternative embodiment of the invention, which issimilar to the first embodiment except that the PWM feedback path hasbeen removed. Instead, the stimulus voltage waveform is applied directlyto the deadband generator and in turn to the complementary FET pair thatexcite the resonance. The removal of the PWM feedback does reduce thelevel of feedback, however there still remains a feedback mechanismthrough the powering method involving C10 and D6. The performance andoperation of the circuit are described below.

FIG. 12 shows graphs of the transponder current, resonance voltage, andthe resonance voltage zoomed in. These graphs correspond to the graphsfor the first embodiment shown in FIG. 7. The level of variation of theresonance voltage is still low, however it has increased by almost anorder of magnitude over the first embodiment (1.1V vs. 150 mV).

FIG. 13 shows the stimulus waveforms in the neighbourhood of the times300 μs and 500 μs, corresponding to high and low current in thetransponder respectively. These graphs correspond to the graphs for thefirst embodiment shown in FIG. 8. FIG. 13A shows the two stimuluspulses, which are now indistinguishable since the PWM section has beenremoved i.e. the pulse width is not modulated anymore. FIG. 13B showsthe current supplied to the resonance through D6, where the differencebetween the two cases is clear. When the transponder current is high thepulse width increases towards earlier time, increasing the energy inputinto the resonance. The mechanism at work here is now described. Whenthe transponder current is high, dissipating additional energy, thecurrent in the reader antenna drops. The current supply pulse through D6has already been described in terms of a transient current into C10 anda subsequent non-transient supply through the diode, once zero netcurrent has flown. Given the drop in the resonance current, thetransient current into C10 is over at an earlier time, giving more tunefor the supply pulse through D6 before the end of the stimulus pulse. Inthis manner, variations in the resonance amplitude are compensated forwith variations in the input power, thus providing negative feedback.Note that this effect is evident in the first embodiment to a lesserextent, with the current supply pulse widening to both positive andnegative time in response to an increase in transponder current. In thatearlier case the effect is reduced by the efficacy of the PWM feedbackthat keeps the resonance amplitude more constant.

Although the level of feedback has been reduced by the removal of thePWM feedback, the performance of the circuit may still be adequate. Theenvelope variations evident in FIG. 12C are not as sharp as FIG. 7C, aneffect that is particularly clear in the sharp cut off of thetransponder current. The reader response time now appears to becomparable to that of the transponder.

FIG. 14 shows a schematic of the present embodiment with a currentmeasurement circuit added, as for the first embodiment. FIG. 15A showsthe original output voltage of the first embodiment, whereas FIG. 15Bshows the output voltage for the present embodiment. The new outputvoltage is lower amplitude and the transitions have been spread out bythe increased time response of the reader. The waveform is however stilla clear measure of the transponder modulation and may be passed on to alevel switch or ADC for subsequent interpretation.

The reduced feedback embodiment described above illustrates that a rangeof alternative implementations are readily possible. With the highestlevels of feedback, the resonance amplitude is kept very constant andthe reader response time may be shorter than that of the transponder.With reduced levels of feedback the resonance amplitude variationincreases together with the reader response time. This latter case canprovide the advantage of a simpler, lower cost circuit whilst stillmaintaining adequate signal quality of the output voltage. Furthermore,because increased resonance amplitude variation is present, themodulation waveform may additionally be taken directly from the envelopeof this voltage waveform, with the system providing the benefit of highefficiency through the use of a high Q reader coil.

FIG. 16 shows a third embodiment of the reader. This embodiment has thesame resonance circuit and reduced feedback level as the secondembodiment. However, the measurement of the input power is different.Rather than measuring the width of the current supply pulse into theresonance through the schottky diode, the width of the precedingtransient current through capacitor C10 is determined. Because the totalpulse width is kept constant (no PWM feedback) any increase in thecurrent supply pulse width is equal and opposite to the change in thewidth of the preceding transient current. As the transient current flowsout of and back into the capacitor C10, the voltage rises and falls, asshown earlier in FIG. 5A. The maximum value of this voltage waveform isdirectly related to the width of this transient pulse. As such, ameasure of the maximum value of this voltage on C10 is a sensitivemeasure of the input power into the resonance. One advantage of thisapproach is a simpler circuit without the need to generate a voltageramp when previously measuring the width of the current supply pulse.

The peak voltage on C10 is stored on capacitor C4 through diode D1, withR3 providing a decay rate that makes the circuit sensitive to movementsin the peak voltage in both directions. The voltage stored on C10 isthen passed onto three opamp stages that provide high pass filtering toremove the static component of the voltage stored on C10, and also lowpass filtering to remove the ripple at the carrier frequency of 125 kHz.The output voltage at Vout is subsequently passed on to an ADC or leveldetector for interpretation of the digital code.

The embodiments described above are based on the method where twocapacitive paths are employed with a continuously variable duty cycle tomatch a range of possible frequencies. One advantage of this approach isthat the reader may be easily tuned to the same frequency as thetransponder. This is particularly advantageous with the feedbackapproach disclosed here, since the behaviour is simplified when thetransformed modulation impedance is mostly resistive. If the reader andtransponder are mismatched then the transformed impedance also includesreactive components, which can distort the output waveform from the nearsquare waves shown in FIG. 15. In fact the shape of the output waveformwhen the transponder is modulating may be used to tune the reader to thetransponder. This may be carried out with each read cycle oralternatively if only a small number of transponders are required to berecognised then this may be carried out in a pre-calibration step,giving one or more desired frequencies at which to operate the reader.Still another alternative method is to determine the transponderresonant frequency using a frequency sweep. Once tuned to thetransponder resonant frequency, the read range of the system ismaximised.

The feedback method may also be applied to a conventional resonantcircuit rather than the preferred, switched capacitor resonant circuitdescribed above. Such an embodiment would also beneficially employ atuning circuit such that the reader drive frequency matches both thereader resonant frequency and the transponder resonance. Theintroduction of negative feedback to keep the resonance amplitudeconstant would have the same beneficial effect as outlined above. Inparticular a high Q coil may be employed for improved efficiency andpotentially greater read range. Implementation with a conventionalresonance can provide the following advantages that could offset theadditional cost and complexity associated with a separate tuningcircuit:

-   1) The conventional resonant circuit may have reduced distortion,    which may be beneficial in fitting the output of the reader within    regulatory limits, particularly at high power.-   2) A conventional resonant circuit does not require the resonance    FET that controls the variable duty cycle of the embodiment    described above. This can be advantageous when operating at high    power, since this situation could require a FET with an expensive    specification.

The previous discussion has been in terms of an FDX RFID system, howeverthis method may also have application for an HDX system. In the HDXsystem negative feedback is used in a similar manner to the FDXembodiments, for the duration of the communication portion of the readcycle. Here the amplitude of the antenna voltage is kept constant at alow level or even zero. The energy input into the reader antenna inorder to maintain this constant level may yield a modulation signalsubstantially unlimited by high Q rate constraints. Such an approach maybe favourable to the prior art that employs a damping circuit to lowerthe reader antenna Q for the duration of the communication cycle. Thismay be particularly favourable when the reader is required to work withboth FDX and HDX transponders, in which case the circuit complexity maybe reduced by employing the feedback method for both types oftransponder, saving system cost.

FIG. 17 shows a fourth embodiment of the invention. The illustratedapplication is an animal entry control system, more specifically a catflap. The cat flap comprises a door, 8, mounted on a hinge, 10, that hasa lock, 9, controlled by an RFID reader. The reader antenna, 11, whichis the same design as the first embodiment, surrounds the entry doorthrough which the cat, 14, is intended to pass. The position of theembedded transponder, 15, in the cat is also shown.

Such an antenna arrangement achieves a large enclosed area for a givenset of outer dimensions of the product. This large area is beneficial inachieving a good read range through a reduction in the rate of fielddecay with distance from the flap. The reader is powered by twobatteries, 13, and the circuit board of the reader, 12, is mounted inthe assembly such that its plane is essentially horizontal. Thisorientation ensures that the predominantly horizontal field lines inthis location from the reader antenna do not lead to significantenclosed flux. This reduces the chance of complications associated withinduced eddy currents.

The reader is designed to register the presence and identificationnumber of a transponder that has been injected under the skin of thecat. For most of the time the reader is not actively reading thetransponder but is in a low power mode, periodically generating afrequency sweep. Absorption from the frequency sweep is monitored toindicate the presence and resonant frequency of the transponder to beregistered when a cat approaches. Once registered the reader moves intoa higher power mode where a steady state energising signal is generatedin the reader antenna at the resonant frequency of the transponder. Thereader employs negative feedback such that high Q rate limitations maybe avoided, and the low loss reader antenna affords improved efficiencyand read range. There is also a tuning step to refine the energisingfrequency to match the transponder frequency, improving the signalquality in the reader. The transponder identification number is read bythe reader and the door unlocked if it matches an earlier storedreference number. The reader subsequently reverts to the low powerproximity detection mode to conserve battery life.

We have described above an RFID reader of sufficient read range and lowpower to make possible a battery operated cat flap sensing a sub-dermaltransponder injected in a cat. As such the owner's cat may be allowed toenter a premises without allowing other animals entry. This applicationavoids the need for collar-mounted keys and therefore can provide thefollowing advantages:

-   1) The cat is often ‘chipped’ with a sub-dermal RFID transponder to    allow identification if lost or for taking the animal across    national borders. As such a separate key does not need to be    supplied with the cat flap, saving cost.-   2) If the animal does not wear a collar then an external key may not    be attached.-   3) External keys can become detached from the collar, which would    lock the animal out from the home. This is not the case for a    sub-dermal transponder.-   4) Some external keys, such as infra-red keys, require battery power    that runs out over time. For the RFID reader only the cat flap    requires battery power, not the key in the animal.

FIG. 18 shows a fifth embodiment of the invention. This embodimentcomprises an antenna 16, reader 17, and lock 18 that is separablymounted on to a standard pet door 19. This has the advantage that theuser does not need to replace a pet door that is already installed intheir house, rather they may fix (retrofit) the external reader to theexisting door. The external reader is mounted on the inside of the houseand the lock prohibits the flap opening inwards when the desired pet isnot registered; this prohibits unwanted animals from entering the house.The flap is free to open outwards, allowing exit to any animal in thehouse.

The antenna mounts onto or around the perimeter of the pet door or flapand the lock mounts at the base of the pet flap, either on the pet flapor on the door or wall in which the flap is provided. Any convenientmounting technique may be employed, for example, gluing, screwing orbolting. The lock is configured to be retrofitted to an existing petdoor, in particular having an upwardly projecting member whichinterferes with the inward opening of the pet door but which stillallows the pet door to open outwards. The lock is under electroniccontrol so that the upwardly projecting member can be controlled toallow inward opening of the pet door, for example by electromagnetic orother means. This control may, for example, retract the upwardlyprojecting member or permit the upwardly projecting member to hinge nearits base.

The antenna, lock, and reader electronics perform the same functions asdescribed for the third embodiment to read the ID number of a sub-dermalRFID chip in any animal approaching the flap. When the desired animal isregistered the lock is released, allowing entry into the house. Thissystem therefore provides the useful function of the full RFID readerpet door shown in the fourth embodiment (FIG. 17) with the addedconvenience of fitting as an addition to existing pct doors.

Applications of embodiments of the invention are not limited to thosedescribed above; rather these are a small subset of possibleapplications. Applications may be found in all existing RFID areas andalso in new fields made possible by the reduced power and extended readrange afforded.

Examples of potential applications of the technology include; forexample:

-   1) Asset tracking-   2) Access control for people or animals-   3) Identification of people or animals.-   4) Animal feeding control-   5) Automatic vehicle identification-   6) Labelling of products in a retail environment, for example for    theft protection or bill totalling.-   7) Storage of information, for example on a credit card or a    passport.

No doubt many other effective alternatives will occur to the skilledperson. It will be understood that the invention is not limited to thedescribed embodiments and encompasses modifications apparent to thoseskilled in the art lying within the spirit and scope of the claimsappended hereto.

1. An RFID tag reader, the reader comprising: an electromagnetic (EM)field generator for generating an electromagnetic (EM) field formodulation by said tag, said modulation comprising modulated load ofsaid EM field by said tag; a detector system responsive to fluctuationsin strength of said em field at said reader; a negative feedback systemresponsive to said detector system to provide a control signal for saidEM field generator for controlling said EM field generator to reducesaid detected fluctuations; and a demodulator responsive to said controlof said EM field to demodulate said EM field modulation by said tag. 2.A tag reader as claimed in claim 1 configured for simultaneous operationof said EM field generator and said detector system.
 3. A tag reader asclaimed in claim 1 wherein said tag comprises a passive tag, and whereinsaid EM field generator is configured to generate a substantiallycontinuous EM field to power said tag.
 4. A tag reader as defined inclaim 1 wherein at least one of said EM em field generator and saiddetector comprises a resonant circuit including a coil with a Q ofgreater than 50, more preferably greater than
 100. 5. A tag reader asdefined in claim 1 wherein said EM field generator and said detectorsystem share a common coil for generating said EM field and detectingsaid EM field fluctuations.
 6. A tag reader as defined in claim 1wherein said feedback system includes an envelope detector to detect anenvelope of said modulated EM field.
 7. A tag reader as defined in claim1 wherein said feedback system further comprises a pulse generator todrive said EM field generator responsive to said detector system.
 8. Atag reader as defined in claim 7 wherein said pulse generator isconfigured to control a power supply to said EM field generator, andfurther comprising a circuit to provide a demand signal, said demandsignal being responsive to an energy drawn per cycle of said EM fieldfrom said power supply by said EM field generator, and wherein saiddemodulator is configured to demodulate said demand signal.
 9. A tagreader as defined in claim 8 further comprising a control system tocontrol a tuning of said detector system responsive to a quality of saiddemodulated demand signal.
 10. A tag reader as defined in claim 1wherein said EM field generator and said detector system comprise ashared controllable electrical resonator.
 11. (canceled)
 12. (canceled)13. A tag reader as defined in claim 1 wherein said modulated load ofsaid EM field by said tag or transponder comprises modulated absorptionof said EM field by said tag or transponder.
 14. A tag reader as claimedin claim 1 incorporated into an animal flap.
 15. A tag reader as claimedin claim 14 wherein said animal flap further comprises an entry controldevice to inhibit entry of an animal through said animal flap and,responsive to said tag reader detecting an animal bearing a said tag, toallow entry of said tagged animal through said animal flap.
 16. An RFIDtag reader claim 1, configured for retrofitting to an animal flap incombination with an animal entry control device to inhibit entry of ananimal through said animal flap and, responsive to said tag readerdetecting an animal bearing a said tag, to allow entry of said taggedanimal through said animal flap, wherein said tag reader and said animalentry control device form a pet entry control system.
 17. (canceled) 18.An RFID system including a transponder and a reader for the transponder,wherein the reader is configured to use negative feedback to increasethe stability of the amplitude of a resonance in the reader in thepresence of a modulating transponder.
 19. An RFID system as claimed inclaim 18 wherein a level of drive into the resonance of a resonantcircuit is measured to demodulate said transponder modulation.
 20. AnRFID system as claimed in claim 18 wherein the residual voltagevariation of said amplitude is measured to demodulate said transpondermodulation.
 21. An RFID system as claimed in claim 18 wherein the systemis a full-duplex system.
 22. An RFID system as claimed in claim 18wherein the system is a half-duplex system.
 23. An RFID system asclaimed in claim 18 wherein said reader comprises a resonant circuitwith at least two capacitive paths and a FET to control a variable dutycycle of these paths to match the resonance to a range of frequencies.24. An RFID system as claimed in claim 19 configured to use the shape ofsaid demodulated modulation to tune the reader to the transponderfrequency.
 25. A method of remotely interrogating a transponder, themethod comprising: generating an electromagnetic (EM) field formodulation by said transponder, said modulation comprising modulatedload of said em field by said transponder; detecting fluctuations insaid EM field caused by said modulation; applying feedback to said EMfield generating to compensate said detected fluctuations; and detectingsaid applied feedback to demodulate said modulation of said EM field bysaid transponder.
 26. Apparatus for remotely interrogating atransponder, the apparatus comprising: means for generating anelectromagnetic (EM) field for modulation by said transponder, saidmodulation comprising load absorption of said EM field by saidtransponder; means for detecting fluctuations in said EM field caused bysaid modulation; means for applying feedback to said EM field generatingto compensate said detected fluctuations; and means for detecting saidapplied feedback to demodulate said modulation of said EM field by saidtransponder.